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LTC1779 250mA Current Mode Step-Down DC/DC Converter in ThinSOT
FEATURES
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DESCRIPTIO
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High Efficiency: Up to 94% 250mA Output Current Wide VIN Range: 2.5V to 9.8V 550kHz Constant Frequency Operation Burst ModeTM Operation at Light Load Low Dropout: 100% Duty Cycle 0.8V Reference Allows Low Output Voltages 2.5% Reference Accuracy Current Mode Operation for Excellent Line and Load Transient Response Low Quiescent Current: 135A Shutdown Mode Draws Only 8A Supply Current Low Profile (1mm) ThinSOTTM Package
The LTC(R)1779 is a constant frequency current mode stepdown DC/DC converter in a 6-lead ThinSOT package. The part operates with a 2.5V to 9.8V input and can provide up to 250mA of output current. Current mode control provides excellent AC and DC load and line regulation. The device incorporates an accurate undervoltage lockout feature that shuts down the LTC1779 when the input voltage falls below 2V. The LTC1779 boasts a 2.5% output voltage accuracy and consumes only 135A of quiescent current. For applications where efficiency is a prime consideration, the LTC1779 is configured for Burst Mode operation, which enhances efficiency at low output current. To further maximize the life of a battery source, the internal P-channel MOSFET is turned on continuously in dropout (100% duty cycle). In shutdown, the device draws a mere 8A. High constant operating frequency of 550kHz allows the use of a small external inductor. The LTC1779 is available in a low profile (1mm) ThinSOT package.
, LTC and LT are registered trademarks of Linear Technology Corporation. Burst Mode and ThinSOT are trademarks of Linear Technology Corporation.
APPLICATIO S
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1- or 2-Cell Lithium-Ion-Powered Applications Cellular Telephones Wireless Modems Portable Computers Distributed 3.3V, 2.5V or 1.8V Power Systems Scanners
TYPICAL APPLICATIO
C3 0.1F 1 20k 100pF 3 VFB SENSE - 4 ITH/RUN LTC1779 GND VIN SW 6 D1 R1 10
L1 22H
C1 10F 16V
VIN 2.5V TO 9.8V VOUT 2.5V 100mA
100 90 VIN = 3.3V 80
EFFICIENCY (%)
+
5
2
C2 47F 6V
169k
70 VIN = 9.8V 60 50 40 30 0.1 VOUT = 2.5V RSENSE = 10 1 10 100 LOAD CURRENT (mA) 1000
1779 F01b
78.7k
1779 F01a
C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT C2: SANYO POSCAP 6TPA47M D1: IR10BQ015 L1: COILTRONICS UP1B220
Figure 1. LTC1779 High Efficiency 2.5V/100mA Step-Down Converter
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Efficiency vs Load Current
VIN = 6V
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LTC1779
ABSOLUTE
(Note 1)
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RATI GS
PACKAGE/ORDER I FOR ATIO
TOP VIEW ITH/RUN 1 GND 2 VFB 3 6 SW 5 VIN 4 SENSE
-
Input Supply Voltage (VIN).........................- 0.3V to 10V SENSE -, SW Voltages .................. - 0.3V to (VIN + 0.3V) VFB, ITH/RUN Voltages ..............................- 0.3V to 2.4V SW Peak Output Current (< 10s) .......................... 0.5A Storage Ambient Temperature Range ... - 65C to 150C Operating Temperature Range (Note 2) ...-40C to 85C Junction Temperature (Note 3) ............................. 150C Lead Temperature (Soldering, 10 sec).................. 300C
ORDER PART NUMBER LTC1779ES6 S6 PART MARKING LTLP
S6 PACKAGE 6-LEAD PLASTIC SOT-23
TJMAX = 150C, JA = 230C/ W
Consult LTC Marketing for parts specified with wider operating temperature ranges.
The q denotes specifications that apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = 4.2V unless otherwise specified. (Note 2)
PARAMETER Input DC Supply Current Normal Operation Shutdown UVLO Undervoltage Lockout Threshold Shutdown Threshold (at ITH/RUN) Start-Up Current Source Regulated Feedback Voltage Output Voltage Line Regulation Output Voltage Load Regulation VFB Input Current Overvoltage Protect Threshold Overvoltage Protect Hysteresis Overtemperature Protect Threshold Overtemperature Protect Hysteresis Oscillator Frequency RDS(ON) of Internal P-Channel FET Peak Current Sense Voltage VFB = 0.8V VFB = 0V VIN = 4.2V, ISW = 100mA (Note 6) 500 VITH/RUN = 0V (Note 5) 0C to 70C (Note 5) - 40C to 85C 2.5V VIN 9.8V (Note 5) ITH/RUN Sinking 5A (Note 5) ITH/RUN Sourcing 5A (Note 5) (Note 5) Measured at VFB 0.820
q q
ELECTRICAL CHARACTERISTICS
CONDITIONS Typicals at VIN = 4.2V (Note 4) 2.5V VIN 9.8V 2.5V VIN 9.8V, VITH/RUN = 0V VIN < UVLO Threshold VIN Falling VIN Rising
q q
MIN
TYP 135 8 7
MAX 240 22 13 2.5 0.5 0.85 0.820 0.830 3
UNITS A A A V V V A V V mV/V mV/A mV/A
1.60 0.15 0.25 0.780 0.770 -3
2.0 2.1 0.325 0.5 0.800 0.800 0 2.5 2.5 5 0.860 30 170 15 550 100 0.85 120
25 0.895
650 1.4
Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: The LTC1779E is guaranteed to meet performance specifications from 0C to 70C. Specifications over the - 40C to 85C operating temperature range are assured by design, characterization and correlation with statistical process controls. Note 3: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formula: TJ = TA + (PD * JC/W)
Note 4: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. Note 5: The LTC1779 is tested in a feedback loop that servos VFB to the output of the error amplifier. Note 6: Peak current sense voltage is reduced dependent upon duty cycle to a percentage of value as given in Figure 2.
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nA V mV C C kHz kHz mV
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LTC1779 TYPICAL PERFOR A CE CHARACTERISTICS
Reference Voltage vs Temperature
825 820 815 VIN = 4.2V 15 12
NORMALIZED FREQUENCY (%)
VFB VOLTAGE (mV)
805 800 795 790 785 780 775 -55 -35 -15 5 25 45 65 85 105 125 TEMPERATURE (C)
1779 G01
TRIP VOLTAGE (V)
810
Shutdown Threshold vs Temperature
600 560 520 VIN = 4.2V 1.85 1.70 1.55 1.40
ITH/RUN VOLTAGE (mV)
480 440 400 360 320 280 240 200 -55 -35 -15 5 25 45 65 85 105 125 TEMPERATURE (C)
1779 G04
RDS(ON) ()
1.25 1.10 0.95 0.80 0.65 0.50
TA = 125C
RDS(ON) ()
PI FU CTIO S
ITH/RUN (Pin 1): This pin performs two functions. It serves as the error amplifier compensation point as well as the run control input. The current comparator threshold increases with this control voltage. Nominal voltage range for this pin is 0.7V to 1.9V. Forcing this pin below 0.325V causes the device to be shut down. In shutdown all functions are disabled and the internal P-channel MOSFET is turned off. The SW pin will be high impedance. GND (Pin 2): Ground Pin. VFB (Pin 3): Receives the feedback voltage from an external resistive divider across the output. SENSE - (Pin 4): The Negative Input to the Current Comparator. Can be connected to VIN for default minimum peak current of 250mA. Connecting a resistor between SENSE - and VIN specifies a lower peak current. (See Applications Information for specifying resistor value.) VIN (Pin 5): Supply Pin. Must be closely decoupled to GND Pin 2. SW (Pin 6): Switching Node and Drain of Internal P-Channel Power MOSFET. Connects to external inductor and catch diode.
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Normalized Oscillator Frequency vs Temperature
VIN = 4.2V 2.20 2.16 2.12 2.08 2.04 2.00 1.96 1.92 1.88 1.84 5 25 45 65 85 105 125 TEMPERATURE (C)
1779 G02
Undervoltage Lockout Trip Voltage vs Temperature
VIN FALLING
9 6 3 0 -3 -6 -9 -12 -15 -55 -35 -15
1.80 -55 -35 -15
5 25 45 65 85 105 125 TEMPERATURE (C)
1779 G03
RDS(ON) of Internal P-Channel FET vs Input Voltage
1.85 ISW = 100mA SENSE - = VIN
RDS(ON) of Internal P-Channel FET vs Temperature
ISW = 100mA 1.70 SENSE - = V IN 1.55 1.40 1.25 1.10 0.95 0.80 0.65 0.50 VIN = 9.8V VIN = 8.4V VIN = 6V VIN = 2.4V VIN = 4.2V
TA = 25C TA = -55C
0.35 2 3 4 5 7 8 6 INPUT VOLTAGE (V) 9 10
0.35 -55 -35 -15
5 25 45 65 85 105 125 TEMPERATURE (C)
1779 G06
1779 G05
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LTC1779
FU CTIO AL DIAGRA
+
ICMP OVERTEMP DETECT 2 RS1 OSC SLOPE COMP R Q S SWITCHING LOGIC AND BLANKING CIRCUIT VIN 1x 24x SW 6 FREQ FOLDBACK SHORT-CIRCUIT DETECT 0.15V VIN EAMP 0.5A
-
+
VIN 0.3V
-
VOLTAGE REFERENCE GND 2 UNDERVOLTAGE LOCKOUT VREF 0.8V
OPERATIO
(Refer to Functional Diagram)
Main Control Loop The LTC1779 is a constant frequency current mode switching regulator. During normal operation, the internal P-channel power MOSFET is turned on each cycle when the oscillator sets the RS latch (RS1) and turned off when the current comparator (ICMP) resets the latch. The peak inductor current at which ICMP resets the RS latch is controlled by the voltage on the ITH/RUN pin, which is the output of the error amplifier EAMP. An external resistive divider connected between VOUT and ground allows the EAMP to receive an output feedback voltage VFB. When the
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-
+
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SENSE - 4 VIN 5 0.3V
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+ -
BURST CMP SLEEP
OVP
+ -
VREF + 60mV
+ -
VREF 0.8V VFB 3 VIN
1 ITH/RUN
0.325V
+
SHDN CMP SHDN UV
-
1.2V
1779FD
load current increases, it causes a slight decrease in VFB relative to the 0.8V reference, which in turn causes the ITH/RUN voltage to increase until the average inductor current matches the new load current. The main control loop is shut down by pulling the ITH/RUN pin low. Releasing ITH/RUN allows an internal 0.5A current source to charge up the external compensation network. When the ITH/RUN pin reaches 325mV, the main control loop is enabled with the ITH/RUN voltage then pulled up to its zero current level of approximately 0.7V. As the external compensation network continues to charge
LTC1779
OPERATIO
up, the corresponding output current trip level follows, allowing normal operation. Comparator OVP guards against transient overshoots > 7.5% by turning off the internal P-channel power MOSFET and keeping it off until the fault is removed. Burst Mode Operation The LTC1779 enters Burst Mode operation at low load currents. In this mode, the peak current of the inductor is set as if VITH/RUN = 1V (at low duty cycles) even though the voltage at the ITH/RUN pin is at a lower value. If the inductor's average current is greater than the load requirement, the voltage at the ITH/RUN pin will drop. When the ITH/RUN voltage goes below 0.85V, the sleep signal goes high, turning off the internal MOSFET. The sleep signal goes low when the ITH/RUN voltage goes above 0.925V and the LTC1779 resumes normal operation. The next oscillator cycle will turn the internal MOSFET on and the switching cycle repeats. Dropout Operation When the input supply voltage decreases towards the output voltage, the rate of change of inductor current during the ON cycle decreases. This reduction means that the internal P-channel MOSFET will remain on for more than one oscillator cycle since the inductor current has not ramped up to the threshold set by EAMP. Further reduction in input supply voltage will eventually cause the P-channel MOSFET to be turned on 100%, i.e., DC. The output voltage will then be determined by the input voltage minus the voltage drop across the MOSFET, the sense resistor and the inductor. Undervoltage Lockout To prevent operation of the P-channel MOSFET below safe input voltage levels, an undervoltage lockout is incorporated into the LTC1779. When the input supply voltage drops below approximately 2.0V, the P-channel MOSFET and all circuitry is turned off except the undervoltage block, which draws only several microamperes.
SF = IOUT/IOUT(MAX) (%)
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(Refer to Functional Diagram)
Short-Circuit Protection When the output is shorted to ground, the frequency of the oscillator will be reduced to about 100kHz. This lower frequency allows the inductor current to safely discharge, thereby preventing current runaway. The oscillator's frequency will gradually increase to its designed rate when the feedback voltage again approaches 0.8V. Overvoltage Protection As a further protection, the overvoltage comparator in the LTC1779 will turn the internal MOSFET off when the feedback voltage has risen 7.5% above the reference voltage of 0.8V. This comparator has a typical hysteresis of 30mV. Slope Compensation and Inductor's Peak Current The inductor's peak current is determined by:
IPK =
M( VITH/RUN - 0.7) 10 RSENSE + 2
(
)
when the LTC1779 is operating below 40% duty cycle. However, once the duty cycle exceeds 40%, slope compensation begins and effectively reduces the peak inductor current. The amount of reduction is given by the curves in Figure 2.
110 100 90 80 70 60 50 40 30 20 10 0 10 20 30 40 50 60 70 80 90 100 DUTY CYCLE (%)
1779 F02
IRIPPLE = 0.4IPK AT 5% DUTY CYCLE IRIPPLE = 0.2IPK AT 5% DUTY CYCLE VIN = 4.2V
Figure 2. Maximum Output Current vs Duty Cycle
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LTC1779
OPERATIO
The variable M is the ratio of the total switch current to that portion of the switch current that flows through RSENSE. M is a function of both RSENSE and ROUT of the internal power switch, which in turn, is a strong function of supply voltage. For values of M refer to Figure 3. In order to guarantee the desired IPK over the full range of supply voltage, the minimum value of M, corresponding to the minimum supply voltage seen in the application, should be chosen. Note that the selection of RSENSE, and hence the resulting M, is an iterative process. For most applications, a value of RSENSE between 0 and 20 will be chosen.
M (mA/mA)
APPLICATIO S I FOR ATIO
The basic LTC1779 application circuit is shown in Figure 1. External component selection is driven by the load requirement and begins with the selection of L1 and RSENSE (= R1). Next, the output diode D1 is selected followed by CIN (= C1)and COUT(= C2). Inductor Value Calculation The inductance value has a direct effect on ripple current. The ripple current, IRIPPLE, decreases with higher inductance or frequency and increases with higher VIN. The inductor's peak-to-peak ripple current is given by:
IRIPPLE =
( VIN - VOUT )VOUT
VIN * * L
where f is the operating frequency fixed at 550kHz in the LTC1779. A smaller value of L results in higher current ripple and output voltage ripple as well as greater core losses. Larger values of L decrease the ripple, but require finding physically larger inductors since maximum DC current rating decreases significantly as inductance increases within inductor product types. Generally, by choosing the desired ripple current based on the maximum output current, the inductor value can be calculated from the previous equation. It is typical to choose the inductor so that the ripple current is about 40% of the maximum output
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(Refer to Functional Diagram)
60 RSENSE = 18.2 55 50 45 40 35 30 25 20 0 1 2 345678 SUPPLY VOLTAGE (V) 9 10 RSENSE = 14 RSENSE = 10 RSENSE = 6.2 RSENSE = 2 RSENSE = 0
1779 F03
Figure 3. M vs Supply Voltage
current at maximum input voltage. Use the following equations to calculate L:
IRIPPLE = 0.4 * IOUT(MAX) L=
(VIN(MAX) - VOUT ) * VOUT
VIN(MAX) * * IRIPPLE IRIPPLE 2
IL(MAX) = IOUT(MAX) +
and then choose an appropriate L and recalculate the ripple current. In Burst Mode operation on the LTC1779, the ripple current is normally set such that the inductor current is continuous during the burst periods. Therefore, the peakto-peak ripple current must not exceed:
IRIPPLE
M(0.030) (RSENSE + 2)
This implies a minimum inductance of:
LMIN = VOUT + VD M(0.030) VIN + VD f (RSENSE + 2) VIN - VOUT
(Use VIN(MAX) = VIN)
LTC1779
APPLICATIO S I FOR ATIO
A smaller value than L MIN could be used in the circuit; however, the inductor current will not be continuous during burst periods. RSENSE Selection for Output Current The selection of RSENSE determines the output current limit, the maximum possible output current before the internal current limit threshold is reached. IOUT(MAX), the maximum specified output current in a design, must be less than ICL. With the current comparator monitoring the voltage developed across RSENSE, the threshold of the comparator determines the inductor's peak current. The maximum output current, ICL, the LTC1779 can provide is given by:
SF 0.12V IRIPPLE ICL = M - 100 RSENSE + 2 2
where SF and M are as defined in the previous section, Figures 2 and 3. Typically, RSENSE is chosen between 0 and 20. Current limit is at a minimum at minimum input voltage and maximum at maximum input voltage. Both conditions should be considered in a design where current limit is important. To calculate several current limit conditions and choose the best sense resistor for your design, first use minimum input voltage. Calculate the duty cycle at minimum input voltage.
DC = VOUT VIN(MIN)
Choose the slope factor, SF, from Figure 2 based on the duty cycle. The ripple current calculated at minimum input voltage and the chosen L should be used in the current limit equation (see Inductor Value Calculation). Figure 3 provides several values of RSENSE and their corresponding M values at different input voltages. Select the minimum input voltage and calculate the resulting minimum current limit settings. The process must be repeated for maximum current limit using duty cycle, slope factor, ripple current and mirror ratio based on maximum input voltage in order to choose the best sense resistor for a particular design and to
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understand how it is going to work over the entire input voltage range. Inductor Core Selection Once the value for L is known, the type of inductor must be selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores, forcing the use of more expensive ferrite, molypermalloy or Kool Mu(R) cores. Actual core loss is independent of core size for a fixed inductor value, but it is very dependent on inductance selected. As inductance increases, core losses go down. Unfortunately, increased inductance requires more turns of wire and therefore copper losses will increase. Ferrite designs have very low core losses and are preferred at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite core material saturates "hard," which means that inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in inductor ripple current and consequent output voltage ripple. Do not allow the core to saturate! Molypermalloy (from Magnetics, Inc.) is a very good, low loss core material for toroids, but it is more expensive than ferrite. A reasonable compromise from the same manufacturer is Kool Mu. Toroids are very space efficient, especially when you can use several layers of wire. Because they generally lack a bobbin, mounting is more difficult. However, new designs for surface mount that do not increase the height significantly are available. Output Diode Selection The catch diode carries load current during the off-time. The average diode current is therefore dependent on the internal P-channel switch duty cycle. At high input voltages the diode conducts most of the time. As VIN approaches VOUT the diode conducts only a small fraction of the time. The most stressful condition for the diode is when the output is short-circuited. Under this condition the diode must safely handle IPK at close to 100% duty cycle. Therefore, it is important to adequately specify the diode peak current and average power dissipation so as not to exceed the diode ratings.
Kool Mu is a registered trademark of Magnetics, Inc.
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LTC1779
APPLICATIO S I FOR ATIO
V -V ID = IN OUT IOUT VIN + VD
Under normal load conditions, the average current conducted by the diode is:
The allowable forward voltage drop in the diode is calculated from the maximum short-circuit current as:
VF
PD ICL (MAX)
where PD is the allowable power dissipation and will be determined by efficiency and/or thermal requirements. A fast switching diode must also be used to optimize efficiency. Schottky diodes are a good choice for low forward drop and fast switching times. Remember to keep lead length short and observe proper grounding (see Board Layout Checklist) to avoid ringing and increased dissipation. CIN and COUT Selection In continuous mode, the source current of the internal P-channel MOSFET is a square wave of duty cycle (VOUT + VD)/(VIN + VD). To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by:
CIN Required IRMS IMAX
[VOUT (VIN - VOUT )]1/ 2
VIN
This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT /2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that capacitor manufacturer's ripple current ratings are often based on 2000 hours of life. This makes it advisable to further derate the capacitor, or to choose a capacitor rated at a higher temperature than required. Several capacitors may be paralleled to meet the size or height requirements in the design. Due to the high operating frequency of the LTC1779, ceramic capacitors can also be used for CIN. Always consult the manufacturer if there is any question.
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The selection of COUT is driven by the required effective series resistance (ESR). Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering. The output ripple (VOUT) is approximated by:
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1 VOUT IRIPPLE ESR + 8 fCOUT
where f is the operating frequency, COUT is the output capacitance and IRIPPLE is the ripple current in the inductor. The output ripple is highest at maximum input voltage since IL increases with input voltage. Manufacturers such as Nichicon, United Chemicon and Sanyo should be considered for high performance throughhole capacitors. The OS-CON semiconductor dielectric capacitor available from Sanyo has the lowest ESR (size) product of any aluminum electrolytic at a somewhat higher price. Once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. In surface mount applications, multiple capacitors may have to be paralleled to meet the ESR or RMS current handling requirements of the application. Aluminum electrolytic and dry tantalum capacitors are both available in surface mount configurations. In the case of tantalum, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS, AVX TPSV and KEMET T510 series of surface mount tantalum, available in case heights ranging from 2mm to 4mm. Other capacitor types include Sanyo OS-CON, Nichicon PL series and Panasonic SP. Low Supply Operation Although the LTC1779 can function down to approximately 2.0V, the maximum allowable output current is reduced when VIN decreases below 3V. Figure 3 shows the amount of change as the supply is reduced down to 2V. Also shown in Figure 4 is the effect of VIN on VREF as VIN goes below 2.3V.
LTC1779
APPLICATIO S I FOR ATIO
105 100 95 90 85 80 75 2.0 VREF
NORMALIZED VOLTAGE (%)
VITH
2.2
2.4 2.6 2.8 INPUT VOLTAGE (V)
3.0
1779 F04
Figure 4. Line Regulation of VREF and VITH
Setting Output Voltage The LTC1779 develops a 0.8V reference voltage between the feedback (Pin 3) terminal and ground (see Figure 5). By selecting resistor R1, a constant current is caused to flow through R1 and R2 to set the overall output voltage. The regulated output voltage is determined by:
R2 VOUT = 0.8 1 + R1
For most applications, an 80k resistor is suggested for R1. To prevent stray pickup, locate resistors R1 and R2 close to LTC1779.
VOUT LTC1779 3 VFB R2
R1
1779 F05
Figure 5. Setting Output Voltage
Efficiency Considerations The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Efficiency can be expressed as: Efficiency = 100% - (1 + 2 + 3 + ...) where 1, 2, etc. are the individual losses as a percentage of input power.
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Although all dissipative elements in the circuit produce losses, four main sources usually account for most of the losses in LTC1779 circuits: 1) LTC1779 DC bias current, 2) MOSFET gate charge current, 3) I2R losses and 4) voltage drop of the output diode. 1. The VIN current is the DC supply current, given in the electrical characteristics, that excludes MOSFET driver and control currents. VIN current results in a small loss which increases with VIN. 2. MOSFET gate charge current results from switching the gate capacitance of the internal power MOSFET. Each time the MOSFET gate is switched from low to high to low again, a packet of charge dQ moves from VIN to ground. The resulting dQ/dt is a current out of VIN which is typically much larger than the DC supply current. In continuous mode, IGATECHG = f(Qp). 3. I2R losses are predicted from the DC resistances of the internal MOSFET, inductor and current shunt. In continuous mode the average output current flows through L but is "chopped" between the internal P-channel MOSFET in series with RSENSE and the output diode. The MOSFET RDS(ON) plus RSENSE multiplied by duty cycle can be summed with the resistances of L and RSENSE to obtain I2R losses. 4. The output diode is a major source of power loss at high currents and gets worse at high input voltages. The diode loss is calculated by multiplying the forward voltage times the diode duty cycle multiplied by the load current. For example, assuming a duty cycle of 50% with a Schottky diode forward voltage drop of 0.4V, the loss increases from 0.5% to 8% as the load current increases from 0.5A to 2A. 5. Transition losses apply to the internal MOSFET and increase at higher operating frequencies and input voltages. Transition losses can be estimated from: Transition Loss = 2(VIN)2IO(MAX)CRSS(f) Other losses including CIN and COUT ESR dissipative losses, and inductor core losses, generally account for less than 2% total additional loss.
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LTC1779
APPLICATIO S I FOR ATIO
Foldback Current Limiting
As described in the Output Diode Selection, the worstcase dissipation occurs with a short-circuited output when the diode conducts the current limit value almost continuously. To prevent excessive heating in the diode, foldback current limiting can be added to reduce the current in proportion to the severity of the fault. Foldback current limiting is implemented by adding diodes DFB1 and DFB2 between the output and the ITH/RUN pin as shown in Figure 6. In a hard short (VOUT = 0V), the current will be reduced to approximately 50% of the maximum output current.
LTC1779 R2 ITH /RUN VFB R1 DFB2
1779 F06
VOUT
+
DFB1
Figure 6. Foldback Current Limiting
1
ITH/RUN LTC1779 GND
RITH
2
3 CITH
VFB
4 SENSE -
BOLD LINES INDICATE HIGH CURRENT PATHS
Figure 7. LTC1779 Layout Diagram (See PC Board Layout Checklist)
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PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC1779. These items are illustrated graphically in the layout diagram in Figure 7. Check the following in your layout: 1. Large switch currents flow into the input capacitor CIN, the power switch and the Schottky diode D1. The loop formed by these components should be as small as possible. 2. Is the input decoupling capacitor (0.1F) connected closely between VIN (Pin 5) and ground (Pin 2)? 3. Keep the switching node SW away from sensitive small signal nodes. 4. Does the VFB pin connect directly to the feedback resistors? The resistive divider R1 and R2 must be connected between the (+) plate of COUT and signal ground. Locate R1 and R2 close to the VFB pin.
VIN
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CIN SW 6 SW L1 VOUT
VIN
5
+
RS 0.1F D1 COUT
R1
1779 F07
R2
LTC1779
PACKAGE DESCRIPTION
S6 Package 6-Lead Plastic SOT-23
(LTC DWG # 05-08-1634) (LTC DWG # 05-08-1636)
SOT-23 (Original) A A1 A2 L .90 - 1.45 (.035 - .057) .00 - 0.15 (.00 - .006) .90 - 1.30 (.035 - .051) .35 - .55 (.014 - .021)
SOT-23 (ThinSOT) 1.00 MAX (.039 MAX) .01 - .10 (.0004 - .004) .80 - .90 (.031 - .035) .30 - .50 REF (.012 - .019 REF) 2.60 - 3.00 (.102 - .118) 1.50 - 1.75 (.059 - .069) (NOTE 3) PIN ONE ID
.20 (.008) DATUM `A' A A2
L NOTE: 1. CONTROLLING DIMENSION: MILLIMETERS MILLIMETERS 2. DIMENSIONS ARE IN (INCHES)
3. DRAWING NOT TO SCALE 4. DIMENSIONS ARE INCLUSIVE OF PLATING 5. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR 6. MOLD FLASH SHALL NOT EXCEED .254mm 7. PACKAGE EIAJ REFERENCE IS: SC-74A (EIAJ) FOR ORIGINAL JEDEL MO-193 FOR THIN
Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
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2.80 - 3.10 (.110 - .118) (NOTE 3)
.95 (.037) REF
.25 - .50 (.010 - .020) (6PLCS, NOTE 2)
.09 - .20 (.004 - .008) (NOTE 2)
1.90 (.074) REF
A1
S6 SOT-23 0401
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LTC1779
TYPICAL APPLICATIONS
LTC1779 Minimal Component Count, Single Li-Ion to 1.8V/250mA Step-Down Converter
C1 10F 16V VIN 2.7V TO 4.2V
EFFICIENCY (%)
1 20k 100pF 3
ITH/RUN LTC1779 GND
SW
6
2
VIN SENSE -
5
VFB
4
C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT C2: SANYO POSCAP 6TPA47M D1: IR10BQ015 L1: COILTRONICS UP1B100
RELATED PARTS
PART NUMBER LT 1375/LT1376 LT1616 LTC1624 LTC1625 LTC1627 LT1676/LT1776 LTC1735 LT1767 LTC1771 LTC1772 LTC1773 LTC1877 LTC1878 LTC3400 LTC3401 LTC3402 LTC3404
(R)
DESCRIPTION 1.5A, 500kHz Step-Down Switching Regulators 600mA Step-Down Switching Regulator High Efficiency SO-8 N-Channel Switching Regulator Controller No RSENSE Synchronous Step-Down Regulator Low Voltage, Monolithic Synchronous Step-Down Regulator Wide Input Range Step-Down Switching Regulators 1.5A, 1.4MHz Step-Down DC/DC Converter Ultralow Supply Current Step-Down DC/DC Controller Constant Frequency Current Mode Step-Down DC/DC Controller 95% Efficient Synchronous Step-Down Controller High Efficiency Monolithic Step-Down Regulator High Efficiency Monolithic Step-Down Regulator 1.2MHz Synchronous Step-Up DC/DC Converter in ThinSOT Single Cell, High Current (1A), Micropower, Synchronous 3MHz Step-Up DC/DC Converter Single Cell, High Current (2A), Micropower, Synchronous 3MHz Step-Up DC/DC Converter 1.4MHz High Efficiency, Monolithic Synchronous Step-Down Regulator
TM
Single, High Efficiency, Low Noise Synchronous Switching Controller High Efficiency 5V to 3.3V Conversion at up to 15A Higher Current, 8-Lead MSOP Package 10A IQ, 93% Efficiency, 1.23V VOUT 18V, 2.8V VIN 20V VIN = 2.5V to 9.8V, IOUT Up to 2A, ThinSOT Package 2.65V VIN 8.5V, 0.8V VOUT VIN, Current Mode, 550kHz 550kHz, MS8, VIN Up to 10V, IQ = 10A, IOUT to 600mA at VIN = 5V 550kHz, MS8, VIN Up to 6V, IQ = 10A, IOUT to 600mA at VIN = 3.3V 92% Efficiency, VIN = 0.5V to 6V, VOUT = 2.6V to 5V VIN = 0.5V to 5V, Up to 97% Efficiency Synchronizable Oscillator from 100kHz to 3MHz VIN = 0.7V to 5V, Up to 95% Efficiency Synchronizable Oscillator from 100kHz to 3MHz Up to 95% Efficiency, 100% Duty Cycle, IQ = 10A, VIN = 2.65V to 6V
No RSENSE is a trademark of Linear Technology Corporation.
12
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 q FAX: (408) 434-0507
q
U
Efficiency vs Load Current
100 95 90 85
VOUT 1.8V 250mA
VIN = 2.7V VIN = 3.6V
L1 10H
80 75 70 65
+
D1
VIN = 4.2V
C2 47F 6V
100k
80.6k
60 55 VOUT = 1.8V RSENSE = 0 1 10 100 LOAD CURRENT (mA) 1k
1779 TA01b
1779 TA01a
50 0.1
COMMENTS High Frequency, Small Inductor, High Efficiency 1.4MHz, 4V to 25V Input, ThinSOT Package 8-Pin N-Channel Drive, 3.5V VIN 36V High Efficiency, No Sense Resistor Low Supply Voltage Range: 2.65V to 8V, IOUT = 0.5A 60V Input, 700mA Internal Switches
1779f LT/TP 0701 2K * PRINTED IN USA
www.linear.com
(c) LINEAR TECHNOLOGY CORPORATION 2000

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